Quadriphase systems using QPSK are commonly used because of their simplicity, efficient operation, resulting narrow bandwidth, and noise tolerance. In a basic QPSK system, the resulting modulated signal has four distinct phase states. These phase states are conveyed by di-bits, each di-bit being typically formed of an I bit and a Q bit.
FIG. 1 illustrates an input data stream 10 in a non-return-to-zero (NRZ) format. In a basic QPSK system without differential encoding, this NRZ data stream is converted using well known techniques into an I data stream 12 and a Q data stream 14, with the resulting di-bit symbol rate equal to half that of the incoming NRZ bit rate. In FIG. 1, the I data is shown being based on only the odd numbered NRZ bits, while the Q data is shown being based on only the even numbered NRZ bits. The I data is generally referred to as being in-phase, and the Q data is referred to as being in quadrature phase. The I and Q data, taken as di-bit symbols, fully convey the information in the NRZ data stream.
FIG. 2 illustrates a differential QPSK system, also referred to as a .pi./4-DQPSK system, where serial data a.sub.n is converted to (X.sub.k, Y.sub.k) symbols by a serial/parallel converter 18 and then changed to corresponding signals (I.sub.k, Q.sub.k) by a differential encoder 19. The symbol (X.sub.k, Y.sub.k) corresponds to two consecutive serial bits (e.g., a.sub.n, a.sub.n +1), while conversion from (X.sub.k, Y.sub.k) to (I.sub.k, Q.sub.k) is performed according to equations 1 and 2 below and Table 1. EQU I.sub.k= I.sub.k -1 cos .DELTA..PHI.(X.sub.k, Y.sub.k)!-A.sub.k -1 sin .DELTA..PHI.(X.sub.k, Y.sub.k)! Eq. 1 EQU Q.sub.k= I.sub.k -1 sin .DELTA..PHI.(X.sub.k, Y.sub.k)!+Q.sub.k -1 cos .DELTA..PHI.(X.sub.k, Y.sub.k)! Eq. 2
TABLE 1 ______________________________________ X.sub.k Y.sub.k .DELTA..o slashed. ______________________________________ 1 1 -3.pi./4 0 1 3.pi./4 0 0 .pi./4 1 0 -.pi./4 ______________________________________
The I and Q data is then filtered by Nyquist low pass filters 20 and 21 to remove (or partially remove) inter-symbol interference.
The I data is then multiplied with a cosine wave by multiplier 22. This cosine wave is shifted 90.degree. by shifter 23 to convert the cosine wave to a sine wave, and the Q data is multiplied with this sine wave by multiplier 24. The modulated I and Q data is then summed by an adder 26 to provide the modulated di-bit symbols at an output of the DQPSK modem.
The generation of such modulated I and Q baseband signals is described in the Personal Handy Phone System RCR standard-28, incorporated herein by reference.
This signal may then be up-converted as necessary and filtered by a bandpass filter, then suitably amplified after any additional up-converting for transmission.
FIG. 3 shows a digital modem 32 which may replace the modem of FIG. 2. In FIG. 3, the incoming data stream is combined with an oversampling clock and converted into a parallel address code by a serial-to-parallel converter 34, which may be a shift register. For each new data symbol or clock bit, a different response is addressed in ROM 36 such that the digital output of ROM 36 generally corresponds to the DQPSK output of adder 26 in FIG. 2. Programming such a ROM 36 with the desired response to the incoming data stream is well known. A digital output of ROM 36 is then applied to a digital-to-analog converter 38 to convert the digital signal to an analog signal for subsequent up-conversion, further filtering, and transmission.
Additional details regarding DQPSK modems may be found in the following publications: the paper entitled, "An Intermediate Frequency Modulator Using Direct Digital Synthesis Techniques For Japanese Personal Handy Phone (PHP) And Digital European Cordless Telecommunications (DECT)," by Bjorn Bjerede, et al., pages 467-471, IEEE Vehicular Technology Conference, Stockholm, Sweden, June 1994; the paper entitled "Digital Modulation/Demodulation Techniques For Mobile Radio Communications In Japan," by Y. Akaiwa, pages 1503-1511, IEICE Transactions, Vol. E 74, No. 6, June 1991; and the book entitled Digital Communications, by Dr. Kamilo Feher, Chapter 4.7.1, Practice-Hall, 1983. All these publications are incorporated herein by reference.
Digital QPSK modems are desirable since they provide more repeatable modulation accuracies. Such digital modulators may be implemented using the ROM look-up table shown in FIG. 3 or using a software program in combination with a processor for providing real-time conversion rather than addressing the DQPSK response stored in a ROM. Digitally storing a filter response in a ROM for use in a digital DQPSK modem is described in copending application, U.S. Ser. No. 08/436,678, entitled "Equalization Filter Compensating For Distortion In A Surface Acoustic Wave Device," filed May 8, 1995, by Daniel Fugue, Gerard Socci, and Benny Madsen, and in U.S. Pat. No. 5,379,242, entitled "ROM Filter," by Dennis Rose and Daniel Fague, both documents being assigned to the present assignee and incorporated herein by reference.
Although using a ROM to generate the filtered DQPSK digital signal has advantages, the resulting die size is relatively large, especially when the data is oversampled. For example, assume that the various possible modulated signals which can occur are stored in the ROM 36 in FIG. 3, the sampling rate is 9.6 MHz, and the symbol rate of the system is 192 Kbps. The carrier frequency of the cosine and sine waves for modulating the I and Q data is set for 1.152 MHz. To ensure that the modulated signal has continuous transitions, an integer number of cycles of the carrier must be stored in the sine and cosine ROMs, whether or not the ROMs are combined with the baseband filter ROMs. Obtaining an integer number of cycles of the carrier can be assured by the following relationship: EQU N/M=f.sub.SAMP /f.sub.CARRIER, Eq.3
where N equals the number of samples of the carrier per symbol, M is the number of complete cycles of the carrier sampled, f.sub.SAMP is the sampling frequency, and f.sub.CARRIER is the carrier frequency. Given the above assumed values, equation 3 gives an N of 25 and an M of 3 as the minimum integers possible. If the required response values were all stored in a ROM, this would mean that each combination of baseband symbols would require 25 samples. The ROM and its addressing logic would have to be run at the sampling frequency 9.6 MHz, causing a large current consumption. An alternate implementation could be to clock a sine and cosine ROM separately from the baseband filter ROMs and use conventional digital multipliers to do the modulation. However, this still has the disadvantage that the sine and cosine ROMs need 25 samples per symbol each.
What is needed is a new implementation of a digital modem which takes up less real estate and consumes less current than the digital modems described above.